Series terminated CMOS output driver with impedance calibration

ABSTRACT

A differential line driver includes a plurality of driver cells. Control logic outputs positive and negative control signals to the driver cells so as to match a combined output impedance of the driver cells at (Vop, Von). Each driver cell includes an input Vip and an input Vin, an output Vop and an output Von, a first PMOS transistor and a first NMOS transistor having gates driven by the input Vip, and a second PMOS transistor and a second NMOS transistor having gates driven by the input Vin. A source of the first PMOS transistor is connected to a source of the second PMOS transistor. A source of the first NMOS transistor is connected to a source of the second NMOS transistor. First and second resistors are connected in series between the first PMOS transistor and the first NMOS transistor, and connected together at Von. Third and fourth resistors are connected in series between the second PMOS transistor and the second NMOS transistor, and connected together at Vop. A first output switch is driven by a corresponding positive control signal and connected between a supply voltage and the sources of the first and second PMOS transistors. A second output switch driven by a corresponding negative control signal and connected between a ground and the sources of the first and second PMOS transistors.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.10/862,467, filed on Jun. 8, 2004, now U.S. Pat. No. 6,894,543, entitledSeries Terminated CMOS Output Driver with Impedance Calibration, whichis a continuation of U.S. patent application Ser. No. 10/419,886, filedon Apr. 22, 2003, entitled Series Terminated CMOS Output Driver withImpedance Calibration, now U.S. Pat. No. 6,771,097, which isincorporated herein by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to line drivers, and more particularly, todifferential line drivers with impedance that is matched to the line.

2. Description of the Related Art

In high speed signaling systems, drivers with output impedance matchedto the line impedance are used. A conventional voltage mode seriesterminated driver can be realized using an inverter with proper sizingof the NMOS and PMOS transistors to give an output impedance equal tothe line impedance, which is typically 50 ohms. A circuit implementationof such a driver 100 in its differential form is shown in FIG. 1.

Referring to FIG. 1, PMOS transistors MP1, MP2 and NMOS transistors MN1,MN2 are sized to give a nominal output impedance of 50 ohms to drive a50 ohm line. To avoid excessive reflections of reverse travelling wavescaused by mismatched far end terminations or discontinuities requiresthat the output impedance of the transmit driver be closely matched tothe characteristic impedance of the line, typically within 10% orbetter. However, the turn-on resistance of the NMOS or PMOS transistorsmay vary by 50% or more across process, supply voltage and temperaturevariations. The driver 100 of FIG. 1 can be improved if the outputimpedance is made up of the sum of the resistance of a transistor switchand a resistor R1 as shown in FIG. 2.

The output impedance of the driver 200 of FIG. 2 can be made to bedominated by the resistor R1, especially since the variation of aresistor across process, voltage and temperature variation is usuallymuch less than that of a transistor. However, the variation is oftenstill in the order of 20 to 30%. Achieving adequate output impedancematching requires some form of calibration or compensation mechanism. Asshown in FIG. 3 and FIG. 4, this can be done by digitally trimming theresistance value if the driver 100 of FIG. 1 or the driver 200 of FIG. 2is replaced by a segmented driver where segments are switched in or outby control lines ctlp0 . . . ctlpi and cltn0 . . . ctlni to match theoutput impedance of the driver to the impedance of the line as closelyas possible.

FIG. 3 shows a segmented driver using the driver cell structure 100 ofFIG. 1, and FIG. 4 shows a segmented driver using the driver cellstructure 200 of FIG. 2. Referring to FIG. 3 each segment of the driveris enabled only if ctlpi and ctlni is asserted high. For example ifctlp1 and ctln1 are asserted high, then transistors MP1 and MN1 areenabled to invert the input signal Vip. This corresponds to reducing theoutput impedance, since the turn-on resistances of MP1 and MN1 are addedin parallel with the existing driver total output impedance.

The control lines ctlp and ctln are usually driven by a feedback controlcircuit that compares the output impedance of a replica output driverwith that of an external reference resistor. After each comparison, afinite state machine uses the control lines ctlp and ctln to turn on oroff driver segments to adjust the total driver output impedance to matchand track the impedance of an external reference resistor.

There are several drawbacks to the driver structure of FIG. 4. Thesignal (Vip, Vin) and control (ctlp, ctln) shares the same logic paththrough the logic gates. The logic gates in each driver segmentintroduce additional delay mismatches between the PMOS and NMOStransistor inputs, since the logic required to turn off an NMOStransistor is different from that for a PMOS transistor. This has theeffect of shifting signal transitions away from the desired voltagepoint. Even if the logic were to be made similar, the loading for theoutput of the logic circuits would be different, since typically PMOStransistors are larger than NMOS transistors, and this can causeadditional delay variation and mismatches.

Also, during the midlevel transition of the driver input voltage Vip,both PMOS and NMOS transistors can be turned on. This causes a largetransient shoot through current given by V_(DD)/(Resistance of PMOStransistor+Resistance of NMOS transistor) at that instant.

A calibration circuit that can be used to calibrate the segmented driveris shown in FIG. 5. (Note that FIG. 5 shows the one half of the circuitrequired to perform impedance calibration for the segmented driver ofFIG. 4.) The circuit in FIG. 5 uses a replica of the PMOS side of thesegmented driver (see William J. Dally and John W. Poulton, “DigitalSystems Engineering” Cambridge University Press, pp. 519–521).

The principle of operation of the circuit of FIG. 5 is as follows: anexternal resistor R_(EXT) and a current source 501 is used to generate avoltage reference at node V1. A matched current source 502 sinks currentfrom the PMOS side of a replica segmented driver. The voltage dropacross the PMOS side output impedance will determine the voltage at V2.The state machine will then turn on each driver segment by assertingctlp0 to ctlpi. As each segment is turned on, the voltage at V2 iscompared to that at V1. When a transition is detected, this indicatesthat the voltage at V1 is nearly equal to that of V1. The outputresistance of the PMOS side is thus matched to that of the externalresistor R_(EXT).

This approach also has a number of disadvantages. Two sets of the samecalibration circuitry consisting of the current sources, comparator,state machine logic are needed to calibrate both the PMOS and NMOS sidesof the driver. Also, two external resistors R_(EXT) and extra pads arerequired. Furthermore, for any transistors that are connected to a paddirectly or through a resistor, special layout rules for ESD(electrostatic discharge) protection are necessary. The special layoutrules cause the actual transistor layout to be much larger (4× to 10×)than without the ESD rules. Referring to FIG. 5, transistors MP0 . . .MPi, as well as the transistors used to create the matched currentsources 501, 502, need to have special ESD layout. Additionally, ifmatched current sources 501, 502 and a reference resistor R_(EXT) withthe same impedance as the output driver are used, then the referencecurrent source needs the same amount of current as the current thatflows through the 50 ohm output driver. This current is significant, andis an inefficient use of available current. Alternatively, the referencecurrent source can be made smaller and the reference resistor larger,but this introduces greater matching and scaling errors.

A similar scheme is mentioned in T. Gabara and S. Knauer, “DigitallyAdjustable Resistors in CMOS for High Performance Applications” IEEEJournal of Solid State Circuits, Vol. 27, No. 8, August 1992, pp.1176–1185, which uses as the first branch an external reference resistorin series with a bottom half of a replica driver to perform calibration.Another branch consists of the other upper half of a replica driverusing another replica of the calibrated bottom half as the referenceresistor. This requires extra area, since it requires three halfreplicas that include two bottom halves and one upper half, as well asextra power in each of the two series branches.

Accordingly, what is needed is a line driver that consumes low currentand whose output impedance is closely matched to that of thetransmission line.

SUMMARY OF THE INVENTION

The present invention is directed to a series terminated CMOS outputdriver with impedance calibration that substantially obviates one ormore of the problems and disadvantages of the related art.

There is provided a differential line driver including a plurality ofdriver cells. Control logic outputs positive and negative controlsignals to the driver cells so as to match a combined output impedanceof the driver cells at (Vop, Von). Each driver cell includes an inputVip and an input Vin, an output Vop and an output Von, a first PMOStransistor and a first NMOS transistor having gates driven by the inputVip, and a second PMOS transistor and a second NMOS transistor havinggates driven by the input Vin. A source of the first PMOS transistor isconnected to a source of the second PMOS transistor. A source of thefirst NMOS transistor is connected to a source of the second NMOStransistor. First and second resistors are connected in series betweenthe first PMOS transistor and the first NMOS transistor, and connectedtogether at Von. Third and fourth resistors are connected in seriesbetween the second PMOS transistor and the second NMOS transistor, andconnected together at Vop. A first output switch is driven by acorresponding positive control signal and connected between a supplyvoltage and the sources of the first and second PMOS transistors. Asecond output switch driven by a corresponding negative control signaland connected between a ground and the sources of the first and secondPMOS transistors.

In another aspect there is provided a differential line driver includinga plurality of driver cells. A feedback loop controls the driver cellswith selective positive and negative control signals to selected drivercells so as to match a combined output impedance of the driver cells.Each driver cell includes a positive input and a negative input, apositive output and a negative output. A first PMOS transistor and afirst NMOS transistor have gates driven by the positive input. A secondPMOS transistor and a second NMOS transistor have gates driven by thenegative input. A source of the first PMOS transistor is connected to asource of the second PMOS transistor. A source of the first NMOStransistor is connected to a source of the second NMOS transistor. Firstand second resistors are connected in series between the first PMOStransistor and the first NMOS transistor, and connected together at thenegative output. Third and fourth resistors are connected in seriesbetween the second PMOS transistor and the second NMOS transistor, andconnected together at the positive output. A first output switch isdriven by a corresponding positive control signal and connected betweena supply voltage and the sources of the first and second PMOStransistors. A second output switch is driven by a correspondingnegative control signal and connected between a ground and the sourcesof the first and second PMOS transistors.

In another aspect there is provided a differential line driver includinga first plurality of parallel driver circuits receiving a positive inputand outputting a negative output. Each driver circuit comprises, inseries between a supply voltage and a ground: a first switch driven by acorresponding positive control signal, a first PMOS transistor whosegate is driven by the positive input, a first resistor, a secondresistor, a first NMOS transistor whose gate is driven by the positiveinput, and a second switch driven by a corresponding negative controlsignal. The negative output is generated between the first and secondresistors. A second plurality of parallel driver circuits input anegative input and output a positive output. Each driver circuitcomprises, in series between the first switch and the second switch: asecond PMOS transistor whose gate is driven by the negative input, athird resistor, a fourth resistor, and a second NMOS transistor whosegate is driven by the negative input. The positive output is generatedbetween the first and second resistors. Control logic generates thepositive and negative control signals so as to match a combined outputimpedance of the differential line driver to a line.

Additional features and advantages of the invention will be set forth inthe description that follows. Yet further features and advantages willbe apparent to a person skilled in the art based on the description setforth herein or may be learned by practice of the invention. Theadvantages of the invention will be realized and attained by thestructure particularly pointed out in the written description and claimshereof as well as the appended drawings.

It is to be understood that both the foregoing general description andthe following detailed description are exemplary and explanatory and areintended to provide further explanation of the invention as claimed.

BRIEF DESCRIPTION OF THE DRAWINGS/FIGS

The accompanying drawings, which are included to provide a furtherunderstanding of the exemplary embodiments of the invention and areincorporated in and constitute a part of this specification, illustrateembodiments of the invention and together with the description serve toexplain the principles of the invention. In the drawings:

FIG. 1 shows a conventional differential line driver cell.

FIG. 2 shows a modified conventional differential line driver cell.

FIG. 3 shows a plurality of conventional driver cells of FIG. 1 arrangedin parallel.

FIG. 4 shows a plurality of conventional driver cells of FIG. 2 arrangedin parallel.

FIG. 5 shows a conventional differential line driver calibrationcircuit.

FIG. 6 shows a differential line driver cell of the present invention.

FIG. 7 shows a calibration circuit for the differential line driver cellof the present invention.

FIG. 8 shows a number of half cells of the present invention arranged inparallel.

FIG. 9 illustrates the calibration process used in the driver of FIGS. 7and 8.

DETAILED DESCRIPTION OF THE INVENTION

Reference will now be made in detail to the embodiments of the presentinvention, examples of which are illustrated in the accompanyingdrawings.

The proposed invention overcomes the disadvantages outlined for both thedriver structure of FIG. 2, the segmented structure of FIG. 4, itsassociated calibration circuitry of FIG. 5 and the similar schemedescribed in Gabara et al.

The output impedance of prior art inverter type drivers as shown in FIG.1 changes significantly with the output voltage. For the presentinvention, if the resistors and transistors are sized such that theresistors provide the dominant termination impedance, then the outputimpedance of the driver will vary insignificantly with respect to theoutput voltage.

Conventional inverter drivers share the same signal and control path. Inthe present invention, the signal path and control path are separated,allowing the removal of any control logic in the signal path.

The transient shoot through current is reduced compared to theconventional inverter type series termination type drivers.

An output series terminated segmented driver cell structure and itsassociated impedance calibration method and circuitry is describedbelow.

An output driver normally includes a number N of driver cells, orsegments. The number of driver cells switched on (or enabled) isdetermined by the impedance calibration circuitry. In one example, thecalibration circuitry can use comparators, a state machine and feedbackloop to switch on or off the required number of driver cells, to matchand track the impedance of an external resistor.

FIG. 6 shows a diagram of a single driver cell 600. Referring to FIG. 6,the driver cell 600 that is enabled by ctlp and ctln that control seriesPMOS and NMOS switches is shown. As shown in FIG. 6, the differentialline driver cell 600 of the present invention includes a switch, forexample, a PMOS transistor MP3 driven by a control signal ctlp. Thetransistor MP3 is connected to sources of two PMOS transistors MP1 andMP2. Drains of the transistors MP1 and MP2 are connected to resistorsRP1 and RP2, respectively, which are in turn connected to resistors RN1and RN2, respectively. The transistors RN1 and RN2 are connected to twoNMOS transistors MN1 and MN2, whose sources are tied together and to asecond switch (for example, an NMOS transistor NM3) that is controlledby a negative control signal ctln.

FIG. 7 shows a diagram of the calibration circuit of the presentinvention. As shown in FIG. 7, an impedance switchable driver D1typically comprises a plurality of driver cells 600, one of which isillustrated in FIG. 6. The PMOS side and the NMOS side both have theirown calibration feedback loops. One of the feedback loops includes acomparator C1, and a state machine L1 that outputs control signals ctlp0. . . ctlpi. Similarly, the NMOS branch includes a comparator C2 and astate machine L2 that outputs the control signals ctln0 . . . ctlni. Areference ladder may be used to generate reference voltages VR1 and VR2for use by the comparators C1 and C2. The circuit is thereforeessentially a dual closed loop system that forces Vop to track VR1 andVon to track VR2. When Vop is equal to VR1 and Von is equal to VR2 thenthe driver output impedance will be 50 ohms on each side.

When ctlp is asserted low to 0 volts and ctln is asserted high to thesupply voltage, the driver cell 600 of FIG. 6 is enabled. Assuming Vipis pulled low, and Vin pulled high, the output impedance of the driverof FIG. 7 at Von is given by the sum of the impedances of transistorsMP3, MP1 and resistor Rp1. The output impedance at Vop is given by thesum of the impedances R(MN3), R(MN2) of transistors MN3, MN2,respectively and resistor Rn2.

R(Von) when Vip=0=R(MP3)+R(MP1)+R(Rp1)

R(Vop) when Vin=V_(DD)=R(MN3)+R(MN2)+R(Rn2)

Transistors MP3 and MP1 are sized such that their impedances, R(MP3),R(MP1), when combined, are less than 10% of R(Von). This provides arelatively small voltage drop across the transistors MP3 and MP1,keeping them in a linear region during the time when they are turned on.In the linear region, the transistors behave like a linear resistor. Bykeeping the transistors MP3 and MP1 in the linear region, R(Von) can bemade to be independent of output voltage at Von.

FIG. 8 shows the segmented driver D1 in additional detail, showing theoverall arrangement of the driver cells 600 of FIG. 6. Note that onlyone side (the PMOS side) of the segmented driver D1 is shown, to makethe diagram simpler. The outputs of all the driver cells 600 are shortedtogether at Vop, Von, such that the total output impedance of thesegmented driver is R(Von)/N, where N is the number of driver cells 600(assuming all N driver cells 600 are enabled). Note that the inputsignal Vip is shorted to the input of all driver cells 600, ensuringthat each driver cell 600 sees substantially the same input signal andloading conditions.

The segmented driver structure D1 of FIG. 8 is calibrated by connectingit's outputs to an accurate external reference resistor R_(EXT) (seeFIG. 7) and placing it in a feedback loop such that the output impedanceof the driver D1 will track the impedance of the reference resistor.

The voltages in FIG. 7 are as follows:VR1=0.75×V _(DD)  Eq. 1VR2=0.25×V _(DD)  Eq. 2

Assuming the comparator C1 has an infinitely large gain and that thestate machine L1 and averaging logic can be modeled as a linear gainblock with a gain of 1, and that there are an infinite number of drivercells 600, then:

$\begin{matrix}{{{Vop} = {{{VR}\; 1\mspace{14mu}{and}\mspace{14mu}{Von}} = {{VR}\; 2}}}\begin{matrix}{{{Iout}({driver})} = {{( {{Vop} - {Von}} )/100}\mspace{14mu}{ohms}}} \\{= {{( {{{VR}\; 1} - {{VR}\; 2}} )/100}\mspace{256mu}{using}\mspace{14mu}{{Eq}.\; 3}}} \\{{= {{V_{DD}/200}\mspace{259mu}{using}\mspace{14mu}{{Eqs}.\; 1}{\mspace{11mu}\;}{and}\mspace{14mu} 2.}}\mspace{11mu}}\end{matrix}} & {{Eq}.\; 3.}\end{matrix}$ $\begin{matrix}{{R({Vop})} = {{( {V_{DD} - {Vop}} )/{Iout}}\mspace{14mu}{driver}}} \\{= {( {V_{DD} - {0.75 \times V_{DD}}} )/}} \\{( {V_{DD}/200} ) = {50\mspace{14mu}{ohms}}} \\{( {{{using}\mspace{14mu}{{Eqs}.\mspace{14mu} 1}},\;{2\mspace{14mu}{and}\mspace{14mu} 3}} ).}\end{matrix}$

Similarly, R(Von)=Von/Iout=VR2/(V_(DD)/200)=0.25×V_(DD)/(V_(DD)/200)=50ohms.

In an actual implementation, assuming a finite number of driver cells600 are used, then the accuracy of the impedance calibration isdetermined by the number N of driver cells 600 covering an impedancerange and by the coupling factor between Von and Vop. The number N ofdriver cells 600 give the minimum impedance step that the impedance canbe calibrated to (in other words the magnitude of the step quantizationerror):R(Von)=R(Vop of single driver cell)/N  Eq. 4where N is the number of driver cells currently switched on.dR(Von)/dN=−R(Vop of single driver cell)/N ²  Eq. 5

dR(Von)/dN gives the incremental impedance change due to a incrementalchange in N. This corresponds to the minimum step accuracy and for afixed impedance of a single cell 600, this depends on N. For example ifR(Vop of single driver cell 600)=1600 ohms, and 32 number of drivercells are switched on, then dR(Von)/dN=−1600/(32²)=1.56 ohms. Theimpedance step change caused by switching one unit impedance cell isthus 1.56 ohms. The minimum voltage step is 1.56×Iout, where lout is thecurrent flowing through the driver and is given by V_(DD)/200 amps.

Referring again to FIG. 7, there are actually 2 feedback loops operatingin the circuit, and they can oppose each other because of couplingbetween Vop and Von (and vice versa). The first feedback loop tries toforce Vop to equal to VR1, and the second loop forces Von to equal VR2.However when Vop changes, Von is affected as well, and vice versa.

When both feedback loops have settled to where Vop is close to VR1 andVon is close to VR2,Vop=VR1±minimum voltage step±ΔVon×KVon=VR2±minimum voltage step±ΔVop×K

where K is the coupling coefficient, ΔVon represents an incrementalchange in Von, and ΔVop represents an incremental change in Vop.

It can be shown that so long as K<1, then both loops will settle andconverge to give Vop approximately equal to VR1 and Von approximatelyequal to VR2. In one embodiment, the K factor is given by the resistorvoltage division ratio and is calculated to be 1/3 for R1=R3 andR2=2×R1.

The impedance error can be calculated as follows:Rerr=dR(Von)/dN+K*dR(Von)/dN  Eq. 6

where dR(Von)/dN is the minimum impedance step change or incrementalimpedance change caused by switching on or off one driver cell.

Using Eq. 5 and Eq. 6, this is given as:Rerr=1.333×R(Vop of single driver cell)/N ²

The following is an algorithm description of the state machine andaveraging logic as implemented in L1 and L2 of FIG. 7. Referring to FIG.9 and FIG. 7, the algorithm works as follows:

When calibration is first started, a default initial number of drivercells 600 (e.g., N=32) are switched on for both the NMOS and PMOS sideof the driver D1. The impedance of the NMOS branch is R(Von)/32 and ofthe PMOS branch is R(Vop)/32. The comparators C1 and C2 then compare Vopwith VR1 and Von with VR2, respectively. If Vop is greater than VR2,this corresponds to R(Von) being greater than the required impedancevalue. This the algorithm will increase the number of turned on NMOSbranches in driver cells 600. The number of cells 600 turned on for theNMOS branch, referred to as Nn here, is stored in a register. Thisprocess of comparing and incrementing or decrementing Nn value isrepeated another seven times, with each Nn value summed or accumulatedwith the previous Nn values. This is denoted as steps 4 to 6 in FIG. 9.When this process has been repeated more than three times (step 9), theneach averaged Nn value is compared with its previous averaged Nn value.If they are equal, this means that the average of the current eight Nnvalues and the average of the previous eight Nn values are equal, andtherefore the impedance value has converged to a stable value. Thealgorithm then outputs the corresponding averaged Nn value to set theimpedance of the driver. Exactly the same process occurs for the PMOSbranch. When both NMOS and PMOS branches have converged to theirrespective impedance values, the calibration clock is stopped and thelogic signals that calibration has completed.

The proposed calibration circuitry has a number of advantages overconventional art. It is much more area efficient compared to theconventional art circuit as shown in FIG. 5 due to the followingreasons:

Matched current sources 501 and 502 of FIG. 5 are not needed and thissaves significant area because these current sources that are connecteddirectly to or indirectly through the PMOS branch to a bonding pad,require special layout rules for ESD protection. This would typicallyincrease the area required for the current sources to 5 to 10 timestheir normal area. In addition, only one external reference resistor isneeded to calibrate both NMOS and PMOS branches for the inventioncompared to two resistors required for the prior art of FIG. 5. Thecalibration circuit proposed by Gabara et al. uses only one referenceresistor as well, but introduces an additional calibrated NMOS branch(i.e., bottom half of driver) to act as a reference resistor tocalibrate the PMOS branch (i.e., upper half of driver). The presentinvention does not require an additional reference resistor nor anadditional NMOS or PMOS branch, and hence results in smaller area. Theproposed driver can be used as a replica for calibration “as is.”

The calibration circuit described herein also is more current efficientthan the implementation as shown in FIG. 5 and as described in Gabara etal. and Dally et al. This is due to the fact that the reference resistoris configured as the actual resistance load of the output driver. Thisenables the current that flows through the driver to flow through thereference resistor as well, unlike the conventional circuit of FIG. 5,which requires additional current to flow through the external referenceresistor to generate a reference voltage. In addition, two referencevoltage generators and four times the amount of current is needed tocalibrate the upper and bottom half of the driver for the schemedescribed in Gabara et al. This is assuming a calibrated impedance of 50ohms, that the branch with the lower half and 50 ohm reference resistortakes V_(DD)/100 amps, and that the branch with the upper and lower halftakes V_(DD)/100 as well. The embodiment of the present inventiondescribed above takes V_(DD)/200, which is four times more currentefficient. Also, assuming the calibrated impedance is 50 ohms, thescheme, as described in Gabara et al., will consume a total of 4× morecurrent since the total current used in the two branches is2×V_(DD)/(2×50). The present invention only requires V_(DD)/(4×50) amps.

The driver cell structure 600 of the present invention also has a numberof advantages over conventional art. Conventional inverter drivers sharethe same signal and control path. In the present invention, the signalpath and the control path are separated, allowing the removal of anycontrol logic from the signal path. This improves signal delay and slopevariation at the input of different driver cells, and also at the inputof the NMOS and PMOS switches.

Furthermore, the transient shoot through current is reduced compared tothe conventional inverter type series termination type drivers. This isdue to the fact that the termination resistors are in series with thetransistor switches. Therefore when both PMOS and NMOS switches are on,the transient shoot through current, which is given by V_(DD) divided bya larger resistance, is reduced.

Conclusion

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample, and not limitation. It will be apparent to persons skilled inthe relevant art that various changes in form and detail can be madetherein without departing from the spirit and scope of the invention.

The present invention has been described above with the aid offunctional building blocks and method steps illustrating the performanceof specified functions and relationships thereof. The boundaries ofthese functional building blocks and method steps have been arbitrarilydefined herein for the convenience of the description. Alternateboundaries can be defined so long as the specified functions andrelationships thereof are appropriately performed. Also, the order ofmethod steps may be rearranged. Any such alternate boundaries are thuswithin the scope and spirit of the claimed invention. One skilled in theart will recognize that these functional building blocks can beimplemented by discrete components, application specific integratedcircuits, processors executing appropriate software and the like or anycombination thereof. Thus, the breadth and scope of the presentinvention should not be limited by any of the above-described exemplaryembodiments, but should be defined only in accordance with the followingclaims and their equivalents.

1. A line driver comprising: a first plurality of driver half cellshaving a common input, and a common output; and control logic thatgenerates control signals for the driver half cells so as to control acombined output impedance of the driver half cells, wherein each driverhalf cell includes: a first PMOS transistor and a first NMOS transistorhaving gates driven by the input; and first and second resistorsconnected in series between the first PMOS transistor and the first NMOStransistor, and connected together at the output, wherein a signal pathfor the input is different from a signal path for the control signals.2. The line driver of claim 1, further comprising a second plurality ofhalf cells having opposite polarities of transistors and connected withthe first plurality of half cells to provide a differential output. 3.The line driver of claim 2, further comprising an external resistorconnected between the outputs of the first and second pluralities ofhalf cells.
 4. The line driver of claim 1, wherein the control logicincludes a state machine.
 5. The line driver of claim 4, furthercomprising a feedback loop connecting the output and the control logic.6. The line driver of claim 5, further comprising a reference ladderwith reference voltages outputted to the feedback loop for use ingenerating the control signals.
 7. The line driver of claim 6, whereinthe feedback loop comprises a comparator that compares a voltage at theoutput with a reference voltage from the reference ladder, and whereinthe state machine outputs positive control signals based on thecomparison.
 8. The line driver of claim 6, wherein the second feedbackloop includes a comparator that compares a voltage at the output with areference voltage from the reference ladder, and wherein the statemachine outputs negative control signals based on the comparison.